Control of MISO Node

ABSTRACT

A method for generating a radio frequency (RF) signal is disclosed herein. The method includes generating, with a plurality of analog-to-digital converters, analog control signals. A modulated control signal is generated, using a modulator, based on the analog control signals. A biased modulated control signal is generated, using an amplifier, based on the modulated control signal and a first bias control signal. Further, the RF signal is generated, using a multiple-input-single-output (MISO) device, based on the biased modulated control signal and a second bias control signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a continuation of U.S. patent application Ser. No. 13/891,316, filed May 10, 2013, now allowed (Atty. Docket No. 1744.21600A3), which is a continuation of U.S. patent application Ser. No. 12/628,510, filed Dec. 1, 2009, now U.S. Pat. No. 8,461,924, which is a continuation of U.S. patent application Ser. No. 12/142,521, filed Jun. 19, 2008, now U.S. Pat. No. 8,013,675, which claims the benefit of U.S. Provisional Patent Application No. 60/929,239, filed Jun. 19, 2007, and U.S. Provisional Patent Application No. 60/929,584, filed Jul. 3, 2007, which are all incorporated herein by reference in their entireties.

The present application is related to U.S. patent application Ser. No. 11/256,172, filed Oct. 24, 2005, now U.S. Pat. No. 7,184,723 and U.S. patent application Ser. No. 11/508,989, filed Aug. 24, 2006, now U.S. Pat. No. 7,355,470, both of which are incorporated herein by reference in their entireties.

BACKGROUND Field

Embodiments of the present invention are related generally to power amplification, modulation, and transmission.

Background

Outphasing amplification techniques have been proposed for RF amplifier designs. In several aspects, however, existing outphasing techniques are deficient in satisfying complex signal amplification requirements, particularly as defined by wireless communication standards. For example, existing outphasing techniques employ an isolating and/or a combining element when combining constant envelope constituents of a desired output signal. Indeed, it is commonly the case that a power combiner is used to combine the constituent signals. This combining approach, however, typically results in a degradation of output signal power due to insertion loss and limited bandwidth, and, correspondingly, a decrease in power efficiency.

What is needed therefore are power amplification methods and systems that solve the deficiencies of existing power amplifying techniques while maximizing power efficiency and minimizing non-linear distortion. Further, power amplification methods and systems that can be implemented without the limitations of traditional power combining circuitry and techniques are needed.

BRIEF SUMMARY

In an embodiment, a method for generating a radio frequency (RF) signal is disclosed herein. The method includes generating, with a plurality of analog-to-digital converters, analog control signals. A modulated control signal is generated, using a modulator, based on the analog control signals. A biased modulated control signal is generated, using an amplifier, based on the modulated control signal and a first bias control signal. Further, the RF signal is generated, using a multiple-input-single-output (MISO) device, based on the biased modulated control signal and a second bias control signal.

In another embodiment, an apparatus for generating a radio frequency (RF) signal is disclosed herein. The apparatus includes a plurality of analog-to-digital converters (ADCs), a modulator, an amplifier, and a multiple-input-single-output (MISO) device. Each of the ADCs is configured to generate an analog control signal. The modulator is configured to generate a modulated control signal based on the analog control signals from the plurality of ADCs. The amplifier is configured to generate a biased modulated control signal based on the modulated control signal and a first bias control signal. And, the MISO device is configured to generate the RF signal based on the biased modulated control signal and a second bias control signal

Each of the ADCs is configured to receive a control signal and to generate an analog control signal. The modulator is configured to generate a modulated control signal based on the analog control signals from the plurality of ADCs. The amplifier is configured to receive the modulated signal and a first bias control signal and to generate a biased modulated control signal. And, the MISO device is configured to receive the biased modulated control signal and a second bias control signal and to generate the RF signal.

Further embodiments, features, and advantages of the present invention, as well as the structure and operation of the various embodiments of the present invention, are described in detail below with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES

Embodiments of the present invention will be described with reference to the accompanying drawings, wherein generally like reference numbers indicate identical or functionally similar elements. Also, generally, the leftmost digit(s) of the reference numbers identify the drawings in which the associated elements are first introduced.

FIG. 1 illustrates two example lossless combiner topologies.

FIG. 2 illustrates an example MISO topology.

FIG. 3 illustrates a conventional branch amplifier topology with a lossless combiner.

FIG. 4 illustrates an example MISO amplifier topology.

FIG. 5 illustrates a traditional topology with lossless combiner.

FIG. 6 illustrates an example MISO topology.

FIG. 7 illustrates an S-parameter test bench for a lossless T-line (lossless T).

FIG. 8 illustrates an example plot of branch impedance for a lossless T.

FIG. 9 illustrates an example plot of branch phase shift for a lossless T.

FIG. 10 illustrates an S-Parameter Smith Chart for a lossless T.

FIG. 11 illustrates an example plot of the phase shift between combiner inputs for a lossless T.

FIG. 12 illustrates an example plot of gain and phase shift versus frequency for a lossless T.

FIG. 13 illustrates an example S-parameter test bench for a MISO combiner node.

FIG. 14 illustrates an example S-parameter Smith Chart for a MISO combiner node.

FIG. 15 illustrates an example plot of phase shift for a MISO combiner node.

FIG. 16 illustrates a sequence of example Smith Chart plots for a lossless Wilkinson T combiner.

FIG. 17 illustrates a sequence of example Smith Chart plots for a pseudo-MISO combiner node.

FIG. 18 illustrates an example S-parameter test bench for a MISO amplifier using ideal switching elements.

FIG. 19 illustrates a theoretical perfect outphasing transfer characteristic along with a simulated transfer characteristic.

FIG. 20 illustrates the behavior of an example MISO amplifier from an efficiency point of view for a variety of control techniques.

FIG. 21 illustrates an example ADS test bench for an example MISO amplifier with blended control.

FIG. 22 illustrates an example ADS test bench for a modified Chireix combiner.

FIG. 23 illustrates the performance of two example Chireix designs and two example MISO designs.

FIG. 24 illustrates example MISO amplifier efficiency versus output power for various control and biasing techniques.

FIG. 25 illustrates control bias points in an example VPA using an example MISO amplifier.

FIG. 26 illustrates an example efficiency performance plot of the system shown in FIG. 25 for various control techniques.

FIG. 27 illustrates example MISO control and compensation for a ramped dual sideband—suppressed carrier (DSB-SC) waveform.

FIGS. 28 and 29 illustrate actual waveforms from a VPA with an example MISO amplifier of a ramped DSB-SC signal along with the MISO/Driver bias control signal and the MISO/Driver collector current.

FIG. 30 illustrates an example WCDMA signal constellation.

FIG. 31 illustrates an example “starburst” characterization and calibration constellation.

FIG. 32 illustrates a single starburst spoke of the example starburst constellation of FIG. 31.

FIG. 33 illustrates example error surfaces.

FIG. 34 illustrates the relationship between error compensation and control functions for the example of FIG. 27.

FIG. 35 illustrates the relationship between the upper and lower branch control, phase control, and vector reconstruction of signals in complex plane.

FIG. 36 illustrates the interrelationship between various example VPA algorithms and controls.

FIG. 37 illustrates an example RL circuit.

FIG. 38 illustrates the relationship between the current through the inductor and the voltage across the inductor in the example RL circuit of FIG. 37 for a 9.225 MHz carrier rate.

FIG. 39 illustrates the relationship between the current through the inductor and the energy stored in the inductor in the example RL circuit of FIG. 37 for a 9.225 MHz carrier rate.

FIG. 40 illustrates the relationship between the current through the inductor and the voltage across the inductor in the example RL circuit of FIG. 37 for a 1.845 GHz MHz carrier rate.

FIG. 41 illustrates the relationship between the current through the inductor and the energy stored in the inductor in the example RL circuit of FIG. 37 for a 1.845 GHz carrier rate.

FIG. 42 illustrates another example RL circuit.

FIG. 43 illustrates the currents through the switch branches in the example RL circuit of FIG. 42 for 90 degrees of outphasing.

FIG. 44 illustrates the current through the inductor in the example RL circuit of FIG. 42 for 90 degrees of outphasing.

FIG. 45 illustrates the current through the switch branches in the example RL circuit of FIG. 42 for 180 degrees of outphasing.

FIG. 46 illustrates the current through the inductor in the example RL circuit of FIG. 42 for 180 degrees of outphasing.

FIG. 47 illustrates the increase in the ramp amplitude of the inductor current for two outphasing angles in the example circuit of FIG. 42.

FIG. 48 illustrates the reactive power cycled in the inductor versus the outphasing angle in the example circuit of FIG. 42.

FIG. 49 illustrates an example RL circuit.

FIG. 50 illustrates current through the inductor of the example RL circuit of FIG. 49 for various outphasing angles.

FIG. 51 illustrates current pulses through the inductor of the example RL circuit of FIG. 49 for various outphasing angles with switch resistance varied.

FIG. 52 illustrates an example circuit having a load coupled by a capacitor.

FIG. 53 illustrates the currents through the inductor and the load in the example circuit of FIG. 52.

FIG. 54 illustrates the effect of varying the switch source resistances on the currents through the inductor and the load in the example circuit of FIG. 52.

FIG. 55 illustrates an example plot of power output versus outphasing angle for a blended outphasing approach.

FIG. 56 illustrates a histogram associated with a WCDMA waveform.

FIG. 57 illustrates the power output to the load in the example circuit of FIG. 52.

FIG. 58 illustrates the average DC current from battery in the example circuit of FIG. 52

DETAILED DESCRIPTION 1. Overview 2. Combiner Definition

3. Outphasing and “Lossy” versus “Lossless” Combiner Attributes

4. Output Branch Amplifiers and MISO Amplifier 5. Simplified Laplace Transfer Functions 6. “Lossless” T and MISO Simulation 7. MISO Amplifier Simulations and “Lossless” T Efficiency

-   -   (a) Switching MISO design         -   (i) Circuit Performance

8. Lab Results for MISO Amplifiers and VPA System

9. VPA Waveform Processing with Blended Control Functions and Compensation

-   -   (a) Calibration and Compensation

10. Comments on Transient Solutions and a Mathematical Basis for the MISO Node Operation

-   -   (a) R-L MISO Circuit without Load and Variable Duty Cycle     -   (b) Blended Outphasing in the R-L Case Without Load     -   (c) Equations Related to Outphasing with a Load         -   (i) Blended Outphasing Efficiency

11. Summary 1. OVERVIEW

A comparison of MISO (multiple input single output) amplifier architectures employing an innovative Vector Power Amplification (VPA) Technology versus conventional outphasing approaches, such as Linear Amplification using Nonlinear Components (LINC), is provided herein. Description of MISO and VPA architectures are presented in U.S. Pat. Nos. 7,184,723 and 7,355,470, both of which are incorporated by reference in their entireties. This comparison is based in simulation results as well as theory. Some concepts are especially expanded on for purposes of this comparison. In particular, since the combining node is a focus of this genre of power amplifier (PA) and is key for many performance metric comparisons, its operation is illustrated in a detailed manner. In addition, an important theme of the comparison is efficiency and its relationship to the combining node operation as well as certain MISO principles.

In sections 2-6, a comparison between MISO and “Lossless” combiners is provided. Section 7 provides simulations which illustrate the efficiency of “Lossless” T and MISO for various blended control MISO approaches. Section 8 provides actual experimental verification of MISO operation and performance, in support of the results presented in Section 7. Section 9 provides the concept of a unified system approach for optimizing the deployment of MISO. Section 10 provides a mathematical view of MISO operation. A summary is presented in Section 11.

2. COMBINER DEFINITION

Generally, a combiner is a structure which permits the transfer of energy from two or more branches or inputs to an output. In contrast, MISO includes structure which facilitates the efficient transfer of energy from a power source to a load. Note that the power source may possess any spectral content, and that this does not require that MISO transfer power at all from its inputs to its output. Indeed, MISO inputs can be viewed as control points that create a variable impedance at the combining node or that permit steering currents to enter or leave the node. As such, in the case of ideal switches at the input branches, no power is transferred through the input branches. This is in direct contrast to all conventional outphasing systems which depend on efficient transfer of power through the input branches of the combiner.

There are two categories of combiners generally, “lossless” and “lossy.” The industry literature distinguishes between the two categories by noting that lossy combiners provide isolation between the input ports and that lossless combiners do not. In this disclosure, a lossless combiner may or may not possess some form of isolation. In particular, significant time delay is considered a certain type of isolation at certain frequencies.

MISO may be described as lossless. However, as the term “lossless” as used by the industry is often misunderstood or misappropriated, it is useful to provide some insight into conventional combiner properties compared to MISO.

First, it should be noted that lossiness refers to the insertion loss properties of the combiner structure, which combines the branches of an outphasing amplifier to generate the amplifier's output. It is also noted that loss is a function of frequency for most circuits involving non-zero branch impedances. While it may be true that a lossless transmission line (lossless T) has little insertion loss at DC and certain harmonically related frequencies associated with the transmission line branches, it is not lossless at all frequencies for fixed termination impedances. As such, lossless Ts typically possess an operational bandwidth.

Two versions of the classical “lossless” combiner often reported in literature are illustrated in FIG. 1. In particular, the top diagram illustrates a “lossless” Wilkinson combiner and the bottom diagram illustrates a Chireix combiner. In contrast, FIG. 2 illustrates a 2-input MISO topology with load.

Some fundamental differences can be noted between the MISO topology and the classical “lossless” combiner topologies. In particular, differences that relate to impedance, time delay, and frequency response can be noted. Other differences will be apparent to persons skilled in the art.

From one aspect, the MISO topology possesses branch impedances that are substantially equal to zero and thus do not alter phase or amplitude characteristics of the branch currents. This results in no restriction in terms of the frequency response of the topology. In contrast, the “lossless” combiner topologies are in fact lossless at only the design frequency and certain multiples thereof, and thus cause frequency-dependent amplitude as well as phase modifications on the branch currents. In fact, significant phase shifts can be experienced at offsets from the fundamental carrier, which considerably restricts the usable frequency response of the topologies.

From another aspect, the “lossless” combiner topologies utilize reflected wave fronts of transverse-electromagnetic energy (TEM) stored in the transmission lines to constructively or destructively interfere in a precise manner, based on wavelength, to provide time delay isolation between the inputs. In contrast, the MISO topology does not store energy temporally and treats all wavelengths equivalently.

In view of the above, “lossless” combiner topologies have limited bandwidths. While it is possible to design an array of wideband “lossless” combiners, “lossless” combiners can never possess all of the following attributes: small size, low complexity, low cost, and can be implemented in IC technology. However, as will be further shown below, the MISO topology provides a substantial improvement, or is even optimal, with respect to any one of these attributes compared to “lossless” combiner topologies.

3. OUTPHASING AND “LOSSY” VERSUS “LOSSLESS” COMBINER ATTRIBUTES

A variety of amplifier architectures have been analyzed in the literature, such as LINC, to evaluate the tradeoffs between lossless and lossy combiners. In particular, Conradi, Johnston, and McRory, “Evaluation of a Lossless Combiner in a LINC Transmitter,” Proceedings of the 1999 IEEE Canadian Conference on Electrical and Computer Engineering, May 1999, found that a lossy matched combiner (which provides some branch isolation) provides better baseline linearity performance in the overall LINC output response of a LINC architecture than a lossless combiner. However, the evaluation found that the lossless combiner provides better efficiency. This can be explained by the fact that a lossy combiner (e.g., Wilkinson) includes a resistor in its topology, which results in power being dissipated in the combiner, adversely affecting the efficiency of the combiner but providing input branch isolation.

MISO optimizes efficiency. In an embodiment, linearity is achieved by means such as VPA Technology used in conjunction with a MISO amplifier.

In terms of efficiency, lossy combiner architectures are not competitive in today's cellular and mobile wireless market. While other techniques may be employed with lossy combiner architectures to trim efficiency losses in the architecture, these techniques often introduce non-linearities.

In embodiments, MISO designs opt for high efficiency combiner architecture and combine characterization, compensation, and/or calibration to address time variant system non-linearities, which are typically associated with transistor technology. As such, MISO enhances efficiency while dedicating digital and/or analog algorithms to linearize the entire VPA architecture, not just the combiner. Further, the necessary combiner properties are maintained without isolation by employing VPA Technology to adjust the combiner sources that drive the combiner's inputs.

4. OUTPUT BRANCH AMPLIFIERS AND MISO AMPLIFIER

In traditional architectures, the branch amplifiers are treated as individual entities with unique inputs and outputs that can be described in terms of their respective transfer characteristics. Whenever these classical amplifiers are employed with a lossy combiner, analysis is manageable provided that the branch amplifiers are at constant bias. However, if conventional amplifiers are employed with a lossless combiner, then the analysis becomes more complicated at the circuit level if complete transistor models are used. That is why system level analysis and characterization are often preferred.

A MISO amplifier is intricate at the circuit level as well but simpler to analyze than a conventional amplifier using a lossless combiner. In fact, embodiments of the MISO amplifier can be viewed as a single unit with multiple inputs and a single output, with the inputs interacting or not. The single output of the exemplary MISO amplifier is a complicated function of the inputs, at the detailed circuit level. However, system level characterization of the MISO amplifier can be simplified when the combiner is viewed as a variable impedance node, with impedance given as a function of outphasing angle or other MISO state. This variable impedance node collects the sum of multiple branch steering currents whenever a power source is available at the summing node. The combiner node uses several means to efficiently integrate the branch currents into a single branch. For example, outphasing, bias control, power control, amplitude control, and/or impedance control can be used. In an embodiment, in the limiting case, when the MISO amplifier input branches are modulated as nearly ideal switches, the combiner node input waveforms are symmetrically pulse width modulated as a function of the outphasing angle. In an embodiment, the modulated pulses repeat at the carrier rate or at some integrally related submultiples or multiple. Further, the pulses are harmonically rich. Therefore, a frequency selective network is utilized to isolate and match the fundamental or desired harmonic to the load.

According to an embodiment of the present invention, a MISO amplifier integrates all of the features of a switching amplifier with the summing node in a single efficient structure. A traditional LINC amplifier, on the other hand, treats the amplifiers in each branch as separate entities to be optimized prior to the cascade of the power combiner.

Raab et al., “RF and Microwave Power Amplifier and Transmitter Technologies—Part 3,” High Frequency Electronics 2003, provides an overview of classes of amplifiers which can be used with the LINC architecture. Similarly, MISO architecture embodiments can use all of these classes of operation. However, in an embodiment, the highest efficiency is attained whenever the branch amplifiers and the MISO amplifier are operated in switch mode.

Consider the two circuit topologies in FIGS. 3 and 4. FIG. 3 illustrates a conventional branch amplifier topology with a lossless combiner. FIG. 4 illustrates a MISO amplifier topology according to an embodiment of the invention. An important difference between the two topologies relates to isolation between the branches. It is noted, for example, that currents at the combining node interact through bandwidth dependent complex impedances in the lossless combiner topology. In contrast, the MISO amplifier topology does not suffer from this restriction. This difference becomes even more important when switch mode operation is invoked. Indeed, the currents flowing from the combiner node through the MISO amplifier can be virtually instantaneously interactive with their branch switching elements. Traditional topologies, on the other hand, do not permit this instantaneous interaction, with either the switch elements or the load branch. In this sense, the lossless T possesses a form of isolation, known as time isolation, while the MISO amplifier generates steering currents or summing currents which contend for the node without isolation.

Switching transient terms and harmonic content are major considerations for the efficiency of a switching amplifier. If the transients are limited or if the harmonic content is misused, it is not possible for the amplifier to operate efficiently. Indeed, the literature is replete with analyses and examples for Class “E” and Class “F” amplifiers which require from 3 to 10 harmonics along with proper management of the conduction angles for current and voltage to obtain a desired result. It is important therefore to note that harmonics are not altered by the combiner transfer function in a MISO amplifier, according to an embodiment of the present invention.

5. SIMPLIFIED LAPLACE TRANSFER FUNCTIONS

The combiner topologies of FIGS. 3 and 4 can be further analyzed using the circuits of FIGS. 5 and 6. FIG. 5 illustrates a traditional topology with lossless combiner. FIG. 6 illustrates an example MISO topology according to an embodiment of the present invention.

Analyzing the circuit of FIG. 5 in the Laplace domain yields:

V 0  ( s ) V S  ( s ) = ( 1 + e - T M  s )  ( ( C + S )  R L ( C + S )  ( ( M + R L ) + ( M + R L + C + S ) ) )

V_(SU)(s)=V_(SL)(s)e^(−T) ^(M) ^(s) (Upper and Lower Branch Source Transform)

V_(S) ΔV_(SU)

T_(M) Δ Delay difference between Upper and Lower Branch Input Signals related to Modulation and Outphasing Angle.

T_(C) Δ Carrier Cycle Time

_(C) Δ Combiner Branch Impedance (T-line description for “Lossless” Wilkinson or other “Lossless” T.)

_(S) Δ Impedance of Upper or Lower Branch Source assumed to be the same.

In these models, the upper and lower branch sources are identical except for the time delay (phase term) related by T_(M). T_(M) can vary according to the modulation envelope. It is possible to create modulation angles such that T_(M) forces a transfer function which approaches zero. An alternate form of the simplified Laplace Transfer Function of the topology of FIG. 5 can be written as:

V 0  ( s ) V S  ( s ) = e - τ D  s  ( 1 + e - T M  s )  ( ( C ′ + S )  R L ( C ′ + S )  ( ( M + R L ) + ( M + R L + C ′ + S ) ) )

_(C)′Δ Impedance Transformation for the “Lossless” Combiner Bench

τ_(D) Δ Effective Delay through the “Lossless” Combiner Branch

In this form it can be seen that the exponent, τ_(D), for combiner branch delay forces different impedance values at different harmonics of the carrier, which in turn modifies V₀(s)/V_(S)(s) at those harmonics.

The transfer function of the MISO topology of FIG. 6 can be written as:

R L  i L  ( s ) V S = R L s  ( α 1  ( s )  ( pu + R a U a  ( s ) ) - α 0  ( s )  R b U a  ( s ) ) α 1  ( s ) = ( U a  ( s ) R a + U b  ( s ) R b )  ( R b U b  ( s ) + R L + M ) - R b  U a  ( s ) R a  U b  ( s ) α 0  ( s ) = U b  ( s ) R b  ( R b U b  ( s ) + R L + M ) $R_{{SX}_{a}} = \frac{R_{a}}{U_{a}(s)}$ $R_{{SX}_{b}} = \frac{R_{b}}{U_{b}(s)}$ ${U_{a}(s)} = {\sum\limits_{k = 0}^{\infty}{\left( {e^{- {ks}} - e^{{- {s{({k + {1/2}})}}}T_{C}}} \right)\; \frac{1}{s}}}$ ${U_{b}(s)} = {\sum\limits_{k = 0}^{\infty}{\left( {e^{- {ks}} - e^{{- {s{({k + {1/2}})}}}T_{C}}} \right)\; \frac{e^{- {s{(T_{M})}}}}{s}}}$

In the MISO model, a main difference is the fact that rather than voltage sources which combine power via the combiner branches, the power transfer occurs from the battery to the combiner node by the operation of switched resistances on the combiner inputs. These switching cycles generate currents, causing energy storage in

_(pu). The energy is then released to the load, when the switches open. The combiner inputs do not transfer power whenever R_(a)=R_(b)=0,∞.

Although V₀(s)/V_(S)(s) was calculated above, for convenience, it should be apparent based on the teachings herein that I₀(s)/I_(S)(s) can be obtained from the relationships:

${I_{0}(s)} = \frac{V_{0}(s)}{R_{L}}$ I S  ( s ) = V S  ( s ) S  ( s )

T_(M) varies according to the outphasing angle and correspondingly forces a different impedance at the combiner node. This is further examined in section 6 below, which illustrates this phenomenon with S-parameters.

6. “LOSSLESS” T AND MISO SIMULATION

A simulation using ADS (Advanced Design System) reveals the differences between conventional lossless combiner topologies and example MISO topologies from an S-Parameter (Scattering parameter) point of view.

FIG. 7 illustrates an S-parameter test bench for a lossless T, which uses a common compound transmission line. Simulations results using the S-parameter test bench of FIG. 7 are illustrated in FIGS. 8-12.

FIG. 8 illustrates an example plot of branch impedance for a lossless T. In particular, FIG. 8 illustrates the branch impedance associated with S31 and shows that the branch impedance associated with S31 varies wildly over the frequency range of several harmonics. Three (3) harmonics are marked for reference. FIG. 9 illustrates an example plot of branch phase shift for a lossless T. In particular, FIG. 9 illustrates the branch phase shift associated with S31 and shows the significant phase shift at each harmonic. It is noted that the phase shift is not modulo λ and that S31 traces multiple loops around the Smith Chart for 3 harmonics.

FIG. 10 illustrates an S-Parameter Smith Chart for a lossless T. In particular, FIG. 10 illustrates a S-Parameter Smith Chart for S(1, 2), S(1, 1), S(1, 3). FIG. 11 illustrates the phase shift associated with S12, which is the phase shift between the combiner inputs. It is important to note the multiple loop trace for S12 to accommodate 3 harmonics. Also to be noted are the drastic variations in both gain and phase shift associated with S33, as illustrated in FIG. 12.

FIG. 13 illustrates an S-parameter test bench for a MISO combiner node. Simulation results using the S-parameter test bench of FIG. 13 are illustrated in FIGS. 14-15.

FIG. 14 illustrates an example S-parameter Smith Chart for a MISO combiner node. In particular, FIG. 14 illustrates an S-parameter Smith Chart for S12, S13, and S11 (S13=S12). FIG. 15 illustrates an example plot of phase shift for a MISO combiner node. In particular, FIG. 15 illustrates the phase shift associated with S31. As shown, the phase shift is minimal and the impedance is constant over the entire range of frequencies. This also holds true for the combiner path to the output load and for the combiner inputs as well.

Also, from this analysis, note that S12=S21=S31 for the MISO combiner node and that there is no phase shift or time delay in the branches.

FIGS. 16 and 17 illustrate sequences of Smith Chart plots, which show how the insertion S parameters change as a function of outphasing angles for three harmonics. In particular, FIG. 16 illustrates the case of a lossless Wilkinson T combiner. FIG. 17 illustrates the case of a pseudo-MISO combiner node. As shown in FIG. 16, in the case of the lossless Wilkinson T, the insertion delay is considerable at each harmonic. It is noted that the sweep spans 90° of outphasing.

Also noted is the difference between the lossless T and the pseudo-MISO for S12 at the fundamental at 1.84 GHz. Also with respect to the harmonics, the lossless T possesses an alternating origin for the locus of points depending on whether the harmonics are even or odd, in addition to the other apparent differences from the pseudo-MISO case. In contrast, in the MISO case, all harmonics launch or spawn from a common point on the real axis. This is significant for asymmetric pulses in switching devices. Indeed, efficient output PA device designs will possess some level of pulse asymmetry to run in an efficient manner. This asymmetry must be accommodated for practical application particularly as outphasing is applied. This asymmetry is more difficult to manage in the lossless T case, for some switching amplifier topologies.

7. MISO AMPLIFIER SIMULATIONS AND “LOSSLESS” T EFFICIENCY

According to an embodiment, the VPA Architecture utilizes a MISO amplifier to enable efficient operations while maintaining a myriad of other benefits. The manner in which the MISO is implemented often utilizes a combiner node with little or no substantial isolation between the inputs and the driving amplifiers or sources. Nevertheless, appropriate current summing at the node is accomplished by controlling the properties of the source amplifiers through a blend of outphasing, bias control, amplitude control, power supply control, and impedance control. In this manner, suitable transfer functions can be constructed to enable efficient transmitter operation, even with waveforms which possess large PAPR (Peak to Average Power Ratio).

One method of accomplishing the desired MISO operation is by blending bias control along with outphasing. Another method could be a blend of amplitude control as well as outphasing and bias control. It is also possible to combine power supply modulation if so desired. The following sections illustrate MISO operation via simulation in two basic configurations. The first set of simulation results relate to a switching MISO architecture captured in the ADS tool using “ideal” switches. The second set of simulation results provides a comparison between the MISO and more traditional lossless combiners using quasi Class “C” amplifier cores. The focus of the comparisons is efficiency versus power output for various levels of output back off. The basic implementation of back off comes from outphasing for the lossless T combiner examples while the MISO examples use additional degrees of freedom to accomplish a similar function.

As will be further described below, the simulations illustrate that the MISO node enables efficient multi branch amplifier signal processing when combined with VPA Technology.

(a) Switching MISO Design

A simulation was conducted in ADS to illustrate the operation of a MISO amplifier using “ideal” switching elements. FIG. 18 illustrates the topology in ADS.

It is noted that the switches represent switching amplifier elements with virtually infinite gain. Parasitics are zero except for the insertion resistance of the switch, which can be controlled from a minimum of 0.1Ω to a very large value. The minimum and maximum values of resistance and controlling function are fully programmable. The functions are deliberately formulated with dependency on outphasing angle, which is swept for many of the simulations using the variable T_(M), which is the time delay between the two source square wave leading edges.

The sources have rise and fall times which can be defined, as well as frequency amplitude and offset. Simulations were conducted in both transient and harmonic balance test benches.

FIG. 19 illustrates a perfect outphasing transfer characteristic along with a simulated transfer characteristic. The perfect characteristic represents the result:

Power Out=10 log₁₀(A cos(φ/2))²+30 dBm

where φ/2 is ½ the outphasing angle. φ is presented on the horizontal axis. The blue trace is the ideal result or template while the red trace is the simulated circuit response. The ideal trace is normalized to the maximum power out for the simulated circuit example. Note that the equivalent outphasing response of the combiner is very close to the ideal response without detailed calibration. In embodiments, this is accomplished by providing simple blended control functions and accuracy is typically forced by detailed compensation or calibration.

FIG. 20 illustrates the behavior of an example MISO amplifier from an efficiency point of view for a variety of control techniques. It is noted that efficiency is poor with pure outphasing control. This is because the branch driving impedances for the inputs to the combiner are not optimized for efficiency versus P_(out). Indeed, the optimal impedance function is non-linear and time variant and is therefore accounted for in the VPA transfer function which is connected to the MISO amplifier.

The other traces in FIG. 20 illustrate several other controls, including “outphase plus bias” control. In practice, this control blend provides very accurate control for precise reconstruction of waveforms such as EDGE. That is, very good ACPR performance is attained using this technique. In addition, it is usually the case that higher output power is available by employing this type of control.

Another control blend uses power supply control in conjunction with phase and bias.

(i) Circuit Performance

This section provides the results of simulations performed on circuits. The circuits are based on Class “C” amplifiers driven by nearly ideal sources, which can be outphased. ADS was utilized as the simulation tool. FIG. 21 illustrates the ADS test bench for a MISO amplifier with blended control. FIG. 22 illustrates the ADS test bench for a modified Chireix combiner.

FIG. 23 illustrates the performance of two Chireix designs and two MISO designs based on the test benches of FIGS. 21 and 22. The cyan trace corresponds to a MISO amplifier with pure outphasing control. The compensated Chireix and classical Chireix are illustrated in green and blue respectively and provide good efficiency versus output power control. Finally, a blended MISO design response is illustrated in violet, with excellent efficiency over the entire output power control range.

Although not illustrated in these simulations, it is also possible to accentuate the efficiency over a particular range of output power to optimize efficiency for specific waveform statistics. This feature is a fully programmable modification of the basic transfer function and control characteristic equation according to embodiments of the present invention.

8. LAB RESULTS FOR MISO AMPLIFIERS AND VPA SYSTEM

This section provides results of laboratory measurements and experiments. Results are included for the MISO amplifier as well as the entire VPA 2 branch RF signal processing chain. The results illustrate the same trend of performance disclosed in the previous sections directed to simulation results.

FIG. 24 illustrates MISO amplifier efficiency versus output power for various control and biasing techniques, for a CW signal.

In addition to MISO amplifier control techniques, blended control techniques can also be used for other stages of the VPA. Accordingly, it is also possible to control bias points along the upper and lower branch amplifier chain, as illustrated in FIG. 25, for example. This allows for the optimization of the entire VPA system, not just the MISO stage. FIG. 26 illustrates the efficiency performance of the system shown in FIG. 25 for a variety of control techniques. As shown, these techniques include not only MISO control techniques but also control techniques applied at other control points of the VPA system.

Typically, blended techniques using outphasing plus bias control provide very good ACPR performance for complicated waveforms, especially over extended dynamic range. In addition, the blended technique generally permits slightly greater compliant maximum output powers for complicated large PAPR waveforms.

9. VPA WAVEFORM PROCESSING WITH BLENDED CONTROL FUNCTIONS AND COMPENSATION

FIG. 25 illustrates a 2 branch implementation of the VPA architecture with a MISO amplifier. As explained in previous sections, the MISO combining node utilizes a combination of outphasing, bias control, amplitude control, power supply control, and impedance control to accomplish an efficient and accurate transfer characteristic. The blend that is selected for the control algorithm is a function of control sensitivity, waveform performance requirements, efficiency, hardware, firmware complexity, and required operational dynamic range. According to embodiments of the present invention, EDGE, GSM, CDMA2000, WCDMA, OFDM, WLAN, and several other waveforms have been demonstrated on a unified platform over full dynamic range in a monolithic SiGe implementation, using bias and outphasing control.

This section explains some aspects of the bias and amplitude control and provides the basis for understanding compensation associated with the VPA. This system level perspective is related to the MISO operation. Since the MISO amplifier consumes a substantial portion of power of all function blocks in the VPA system it must be operated and controlled correctly in the context of efficient waveform reconstruction. This demands a symbiotic relationship between the MISO amplifier, drivers, pre-drivers, vector modulators, and control algorithms.

FIG. 27 illustrates MISO control and compensation for a ramped dual sideband—suppressed carrier (DSB-SC) waveform, according to an embodiment of the present invention. The plots shown in FIG. 27 are generated using actual lab results.

The top plot in FIG. 27 illustrates the envelope for desired a linearly ramped RF waveform. The envelope represents the magnitude of the baseband waveform prior to algorithmic decomposition. Although not explicitly indicated in this set of plots the RF waveform changes phase 180° at the zero crossing point (sample 200). The three subsequent plots in FIG. 27 show the control functions required to reproduce the desired response at the filtered MISO output. In particular, the second plot is a plot of outphasing angle. The upper trace is a reference plot of the outphasing angle in degrees for a pure “ideal”, outphasing system. The lower trace illustrates an actual outphasing control portion of a blended outphasing plus bias control algorithm. Note that the full outphasing angle is approximately 75°, as opposed to 180° for purely outphased systems.

The third plot in FIG. 27 illustrates bias control, which operates in concert with outphasing control to produce the blended attenuation of the ramp envelope all the way to the MISO output. This control includes compensation or calibration for amplitude mismatches, phase asymmetries, etc. Therefore, the control waveforms provided must take into account the non-ideal behavior of all of the components and functions in the dual branch chain. Both driver and MISO stages are controlled for this example. Controlling the bias of the driver reduces the input drive to the MISO as a function of signal envelope

The fourth lot at the bottom of FIG. 27 illustrates the required phase control to guarantee phase linearity performance of the waveform over its entire dynamic range. At sample 200 a distinct discontinuity is apparent, which is associated with the 180° inversion at the ramp's zero crossing. Note that the phase compensation is very non-linear near the zero crossing at sample 200. At this point, the amplifiers are biased near cut off and S21 is very ill behaved for both the driver and PA/MISO stage. The plot associated with phase compensation represents the additional control required for proper carrier phase shift given the drastic shift in phase due to changing bias conditions. It should be further stated that the relationship between amplitude control and phase is such that they are codependent. As outphasing and bias are adjusted to create the signal envelope the phase compensation requirements change dynamically.

FIGS. 28 and 29 illustrate actual waveforms from a VPA with MISO, according to an embodiment of the present invention, which show the ramped DSB-SC signal along with the MISO/Driver bias control signal and the MISO/Driver collector current.

(a) Calibration and Compensation

Any control algorithm must contemplate the non ideal behavior of the amplifier chain. AM-PM and PM-AM distortion must also be addressed. Many signals of interest which are processed by the VPA architecture can be represented by constellations and constellation trajectories within the complex signaling plane. The RF carrier at the output of the filtered MISO can be written in complex form as:

y(t)=A(t)e ^(j(ω) ^(C) ^(t+Θ))

ω_(C) Δ Carrier Frequency

ΘΔ Modulation Angle

A(t)Δ Amplitude Modulation

The complex signal plane representations of interest are obtained from the Re{y(t)} and Im{y(t)} components. A goal for the VPA is to produce signals at the filtered MISO node such that when decomposed into their complex envelope components, and projected onto the complex signaling plane, the distortions are minimal and efficiency is maximized. An example for such a signal constellation is given in FIG. 30.

Therefore, if the VPA response is known for the entire complex plane, a suitable algorithm will adjust the control signals to contemplate all distortions. If the control is accurate enough given the required compensations, compliant signal reconstruction is possible at the combining node.

Suppose that the VPA system is stimulated in such a way that the output RF signal can be processed to detect the system non-linearities, superimposed on the complex signaling plane. FIG. 31 illustrates an example “starburst” characterization and calibration constellation, which is one technique to characterize the non-linearities imposed by the system as measured at the combining node, after filtering.

The red radials represent stimulus signals in the complex plane. The radials are swept out to the unit circle and back to the origin for a plethora of angles. Each swept radial at the system input spawns a corresponding blue radial at the output. Notice the bifurcation in the blue radials. This arises due to the memory inherent in high power SiGe transistor devices operated over certain portions of their dynamic range. It can be inferred from the complex plane starburst stimulus/response that a significant amount of amplitude and phase non-linearity is present. This non-linearity can be captured as an error (

_(ε) _(R) ) and characterized.

FIG. 32 illustrates a single starburst spoke of the example starburst constellation of FIG. 31. As shown in FIG. 32, for each point along the trajectory, there is a magnitude and phase error or distortion which is related to error vector, D _(ε) _(R) . If the components of the error vector are disassembled into magnitude and phase for the entire complex plane, then error surfaces can be constructed as shown in FIG. 33.

The magnitude error |D_(ε) _(R) | must be taken into account by the bias and outphasing control. The phase compensation must take into consideration the phase error ∠D_(ε) _(R) . FIG. 34 illustrates the relationship between error compensation and control functions for the example of FIG. 27.

Another way to examine the role of the controls in signal reconstruction is to associate the controls with their impact on vector operations in the complex plane. The vectors are outphasing signals generated in the upper and lower branches of the hardware, which drives and includes the MISO amplifier. The outphasing angle is controlled, along with the gain of the driver and gain of the MISO amplifier.

FIG. 35 illustrates the relationship between the upper and lower branch control, phase control, and vector reconstruction of signals in complex plane.

The upper and lower branch signals at the MISO amplifier inputs are controlled in terms of bias and outphasing angle φ to create the appropriate amplitude at the MISO combining node. In addition, the amplitudes of the MISO inputs may be controlled by varying the driver bias. |

_(ε) _(R) | must be taken into account when implementing the blended control.

The phase of the reconstructed carrier is given by Θ and is controlled by manipulation of vector modulators at each branch input or controlling the phase of the RF carrier directly at the RF LO synthesizer for the system (or both). This phase Θ is associated with the modulation angle of the application signal, y(t).

Although a distinction is drawn between amplitude control and phase control of Θ they are in fact dependent and interact considerably. Their respective calculations and compensations are conjoined just as the real and imaginary components of

_(ε) _(R) are interrelated. This is due to AM-PM and PM-AM distortion in physical hardware. As bias is reduced or power supplies varied or impedances tweaked, the insertion phase of both branches of a two branch MISO changes and therefore is a component of ∠

_(ε) _(R) . Hence, generating a particular point or trajectory in the complex plane related to the complex signal, at the filtered MISO output, requires the solution of at least a two dimensional parametric equation. The numbers generated for amplitude control of the complex signal are part of the solution for Θ, the modulation angle, and vice versa.

FIG. 36 illustrates the interrelationship between various example VPA algorithms and controls.

10. COMMENTS ON TRANSIENT SOLUTIONS AND A MATHEMATICAL BASIS FOR THE MISO NODE OPERATION

Prior sections have shown that a purely outphased MISO is very efficient at maximum power output for a zero degree outphasing angle. Efficiency was also shown to drop as outphasing is applied. However, when the MISO amplifier is used in conjunction with VPA principles and control algorithms, improved efficiency is maintained throughout the power range of interest. This section provides a mathematical treatment of this behavior, including differential equation analysis which corroborates the simulated data presented previously.

Transient analysis usually requires the solution of a system of equations. In the continuous or quasi-continuous case, a time domain solution for t=(0⁺→∞) demands the solution of 2^(nd) order differential equations with time variant coefficients. Such equations are not practical to solve in closed form. Nevertheless, some piecewise solutions and formulation of the problem can yield important insights.

(a) R-L MISO Circuit without Load and Variable Duty Cycle

A principle of the switching amplifier requires that the power source transfers energy to some sort of energy storage device and then removes the energy (or allow it to “siphon” off) at some time later by efficient means for use by a load. The energy storage device may be a capacitor, inductor, transmission line, electrically coupled circuit, magnetically coupled circuit, or hybrids of all of the above. The transmission line is often overlooked as a means of storing energy. However, the TEM modes do in fact transport energy and can be efficient, provided loss tangents are minimal. The following treatment illustrates the use of an inductor to store energy. Consider the example RL circuit illustrated in FIG. 37.

Suppose the switch is closed at t=0. At t→0⁺ the characteristic equation of the system may be used to determine the current flowing in the circuit and the energy stored in the inductor. The differential equation is:

${\frac{{Ldi}_{a}}{dt} + {{Rs}_{X_{A}}i_{a}}} = V_{S}$

The solution of this equation requires the consideration of initial conditions:

i_(a) (t=0⁻)=0

i_(a) (t=0⁺)=0

Therefore,

$\frac{{di}_{a}}{dt} = \frac{V_{S}}{L}$

The final current at t→∞ must approach Vs/R_(SX) _(A) .

This yields a solution of:

$i_{a} = {\frac{V_{S}}{R_{{SX}_{A}}}\left( {1 - e^{{- {({R_{{SX}_{A}}/L})}}t}} \right)}$

FIG. 38 illustrates the relationship between the current through the inductor and the voltage across the inductor in the example RL circuit of FIG. 37 for a 9.225 MHz carrier rate.

FIG. 39 illustrates the relationship between the current through the inductor and the energy stored in the inductor in the example RL circuit of FIG. 37 for a 9.225 MHz carrier rate.

The circuit uses a 1 volt battery, 5 nH inductor and 1Ω resistor (R_(SX) _(A) ).

FIGS. 38 and 39 are from a simulation that permits the inductor current to reach a maximum limit. This can be accomplished by controlling the duty cycle and frequency of the switch, such that the switching cycle time is virtually infinite compared to the circuit time constant.

If the switch frequency is required to operate at the carrier rate, then only a fraction of the energy may be stored in the inductor given the same circuit values, if the carrier cycle time is considerably shorter than the time constant (R_(SX) _(A) /L). For instance, consider the carrier rate of 9.225 MHz in the example illustrated in FIGS. 38 and 39 compared to the example illustrated in FIGS. 40 and 41, which uses a switch rate of 1.845 GHz, with the same circuit values.

FIG. 40 illustrates the relationship between the current through the inductor and the voltage across the inductor in the example RL circuit of FIG. 37 for a 1.845 GHz MHz carrier rate.

FIG. 41 illustrates the relationship between the current through the inductor and the energy stored in the inductor in the example RL circuit of FIG. 37 for a 1.845 GHz carrier rate.

It is noted from FIGS. 40 and 41 that the current in this time domain range is approximately a ramp. Also, the average current in the inductor is illustrated in FIG. 40. This is an important consideration for efficiency calculations since the average current through the resistor R_(SX) _(A) causes power to be dissipated. In this example, the resistor dissipates approximately 1.6 mW.

Suppose that the R-L switched circuit of FIG. 37 is augmented with another switch. Further suppose that the two switches can be outphased. The circuit is illustrated in FIG. 42.

Assuming that the time delay between pulses controlling the switches S_(X) _(A) and S_(X) _(B) is given by T_(M), which is related to the outphasing angle, there are 4 equivalent resistance values which are relevant over certain segments of time. These represent 4 unique time constants given by:

$\tau_{1} = \frac{R_{{SX}_{A}}}{L}$ $\tau_{2} = {\left( \frac{R_{{SX}_{A}} \cdot R_{{SX}_{B}}}{R_{{SX}_{A}} + R_{{SX}_{B}}} \right)/L}$ τ₃ = (R_(SX_(B))/L) τ₄ = ∞

Taking into account an infinite pulse train, the following cyclic unit step functions are constructed and associated with each time constant:

$U_{1} = {\sum\limits_{k = 0}^{\infty}\left\lbrack {\left( {u\left( {t - {kT}_{C}} \right)} \right) - {u\left( {t - \left( {{kT}_{C} + T_{M}} \right)} \right)}} \right\rbrack}$ $U_{2} = {\sum\limits_{k = 0}^{\infty}\left\lbrack {\left( {u\left( {t - {kT}_{C} + T_{M}} \right)} \right) - {u\left( {t - \left( {T_{C}\left( {k + {1/2}} \right)} \right)} \right)}} \right\rbrack}$ $U_{3} = {\sum\limits_{k = 0}^{\infty}\left\lbrack \left( {{u\left( {t - {\left( {k + {1/2}} \right)T_{C}}} \right)} - {u\left( {t - \left( {{T_{C}\left( {k + {1/2}} \right)} + T_{M}} \right)} \right)}} \right) \right\rbrack}$ $U_{4} = {\sum\limits_{k = 0}^{\infty}\left\lbrack \left( {u\left( {t - \left( {{T_{C}\left( {k + {1/2}} \right)} + T_{M}} \right) - {ut} - {T_{C}\left( {k + 1} \right)}} \right)} \right) \right\rbrack}$

T_(C) Δ (Carrier Rate)⁻¹

T_(M) Δ Time Delay associated with outphasing modulation angle.

Inspection reveals that u_(v) do not overlap in time. Therefore the differential equation solution will take on the following form for i_(S):

$i_{S} = {\frac{Vs}{L{\sum\limits_{i = 1}^{4}{u_{i}\tau_{i}}}}\left( {1 - e^{- {\sum\limits_{i = 1}^{4}{u_{i}\tau_{i}t}}}} \right)}$

Currents through R_(SX) _(A) and R_(SX) _(B) illustrate the multiple time constant when superimposed on the same graph. The switching events are illustrated along with the switch currents in FIG. 43, which illustrates the currents through the switch branches in the example RL circuit of FIG. 42 for 90 degrees of outphasing.

τ₁, τ₂, τ₃, τ₄ are visible for this 90 degree outphase example. The current through the inductor is the sum of both switch branches and is illustrated in FIG. 44.

At the 180 degrees of outphasing, the power source is continually shunted through the switches to ground. This condition eliminates all harmonic currents in the inductor. Only DC currents remain, so that the fundamental is eliminated. This is illustrated in FIGS. 45 and 46. FIG. 45 illustrates the current through the switch branches in the example RL circuit of FIG. 42 for 180 degrees of outphasing. FIG. 46 illustrates the current through the inductor in the example RL circuit of FIG. 42 for 180 degrees of outphasing.

Whenever the outphasing angle <180° the current through the pull up inductor is a ramp:

$i_{A} = {\frac{1}{L}{\int{V_{L}{dt}}}}$

Therefore, when the switches are closed a voltage is applied across the inductor and a current ramp is generated. The outphasing angle has an effect of extending the duty cycle of the applied voltage from T_(C)/2 up to a full duty cycle, T_(C). As illustrated above at the full duty cycle, T_(C), the switches are completely outphascd and the ramp disappears.

Three observations are gleaned from this discussion thus far: a) The current flowing through the energy storage element (inductor) is approximately a linear ramp and extends in duty cycle from T_(C)/2 to T_(C)−; b) The current ramp will grow to twice the magnitude as the duty cycle increase, in a linear manner; and c) The average current flowing in the inductor and the resistors increases as the duty cycle of the ramp increases up to the 180° outphasing angle, at which time only DC currents flow and are equal to a value of the battery voltage divided by the switch resistance.

a), b), and c) along with the prior differential equation analysis can be used to examine “purely outphased” MISO, particularly when efficiency versus power out is examined.

The circuit analyzed above does not possess a real load. Nevertheless, a pseudo-efficiency can be defined as follows:

$\eta_{eff} \approx \frac{{Reactive}\mspace{14mu} {Power}\mspace{14mu} {in}\mspace{14mu} {the}\mspace{14mu} {Fundamental}\mspace{14mu} {Harmonic}}{{Total}\mspace{20mu} {DC}\mspace{14mu} {Power}\mspace{14mu} {Supplied}\mspace{14mu} {to}\mspace{14mu} {the}\mspace{14mu} {Circuit}}$

The Fourier coefficients for the general current ramp with variable duty cycle can be written as:

$a_{n} = {\sum\limits_{n = 1}^{\infty}\left\lbrack {{\frac{A}{\left( {n\; \pi} \right)^{2}}\left( {{\cos \left( {\left( {\frac{1}{2} + \frac{T_{M}}{T_{C}}} \right)2n\; \pi} \right)} - 1} \right)} + {\frac{2A}{n\; \pi}{\sin \left( {\left( {\frac{1}{2} + \frac{T_{M}}{T_{C}}} \right)2n\; \pi} \right)}}} \right\rbrack}$ $\mspace{20mu} {b_{n} = {\sum\limits_{n = 1}^{\infty}\left\lbrack {\frac{A}{\left( {n\; \pi} \right)^{2}} + {\frac{2\left( {\frac{1}{2} + \frac{T_{M}}{T_{C}}} \right)A}{n\; \pi}\left( {1 - {\cos \left( {\left( {\frac{1}{2} + \frac{T_{M}}{T_{C}}} \right)2n\; \pi} \right)}} \right)}} \right\rbrack}}$

where

T_(M) Δ Time Delay associated with outphasing modulation

${\varphi \; \underset{\_}{\Delta}\mspace{14mu} {Outphasing}\mspace{14mu} {Angle}} = {{\left( \frac{2T_{M}}{T_{C}} \right) \cdot 180}\mspace{14mu} {degrees}}$

Reactive power is calculated in the fundamental (n=1) and is found to increase as the duty cycle increased for the ramp. This can be explained by the ramp amplitude almost doubling, near 180 degrees outphasing. Although the harmonic energy due to the duty cycle decreases slightly, the increase in ramp amplitude over the cycle duration overcomes the Fourier losses. FIG. 47 illustrates the increase in the ramp amplitude of the inductor current for two outphasing angles in the example circuit of FIG. 42.

The reactive power cycled in the inductor is plotted in FIG. 48 versus φ with power in dB_(r) using φ=0° as the reference of zero.

In addition, the pseudo efficiency is given as a function of outphasing angle relative to the efficiency at φ=0°. Notice that pseudo efficiency is flat up to more than 100° of outphasing. This is further discussed below.

For a range of 0 degrees outphasing to nearly 180 degrees of outphasing, the reactive power in the fundamental cycled through the inductor actually increases. In other words, pure outphasing with a MISO node, and nearly ideal switching components, does not produce an attenuation of the available output energy over virtually the entire outphasing angle range. Furthermore, as the current ramp increases in duty cycle from 50% to nearly 100%, the DC average increases by 6 dB until just prior to the 100% duty cycle. Again, this is extracted from the Fourier analysis of the ramped current, revealed by the differential equation.

Although a load circuit is not included for this basic analysis, energy available in the inductor can be transferred to a load.

(b) Blended Outphasing in the R-L Case without Load

In the prior section, it was verified by mathematical analysis that the reactive energy transferred to the inductor from the battery at the fundamental frequency does not fall off rapidly as a function of outphasing angle. In classical outphasing systems, the power available would normally follow a characteristic of:

P _(OUT) ∝A ² cos²(φ/2)

In the MISO case, pure outphasing does not produce this result. Rather, the energy available in the inductor increases as was illustrated by the differential equation for the simple MISO example.

In this section, the simple example is revisited with the goal of illustrating how source impedances (more specifically resistances for this simple example) modify the energy transfer characteristic.

The appropriate differential equation is derived from example circuit shown in FIG. 49.

The relevant circuit equations are:

${V_{S} - {L\frac{{di}_{a}}{dt}} + {\left( {i_{b} - i_{a}} \right)\frac{R_{a}}{\sum\limits_{k = 0}^{\infty}{u\left\lbrack \underset{\underset{u_{a}}{}}{\left( {t - {kT}_{C}} \right) - {u\left( {t - {\left( {k + {1/2}} \right)T_{C}}} \right)}} \right\rbrack}}}} = 0$ ${{\left( {{- i_{b}} + i_{a}} \right)\frac{R_{a}}{\sum\limits_{k = 0}^{\infty}\left\lbrack {{u\left( {t - {kT}_{zc}} \right)} - {u\left( {t - {\left( {k + {1/2}} \right)T_{C}}} \right)}} \right\rbrack}} - {i_{b}\frac{R_{b}}{\sum\limits_{k = 0}^{\infty}\left\lbrack \underset{\underset{u_{b}}{}}{u\left( {t - \left( {{kT}_{C} + T_{M}} \right) - {u\left( {t - {\left( {k + {1/2}} \right)T} + T_{M}} \right)}} \right)} \right\rbrack}}} = 0$

Two system equations result:

${\left( {\frac{d}{dt} - \frac{1}{L\left( {\frac{u_{b}}{R_{b}} + \frac{u_{a}}{R_{a}}} \right)}} \right)\; i_{b}} = {\frac{1u_{b}}{{LR}_{b}}V_{S}}$ $i_{a} = {\left( {1 + \frac{R_{b}u_{a}}{R_{a}u_{b}}} \right)\mspace{11mu} i_{b}}$

Hence, the general solution can be written as:

${i_{b} = {k_{b}\left( {1 - e^{{- \underset{\underset{\lambda}{}}{(\frac{\frac{R_{a}}{U_{a}} + \frac{R_{b}}{U_{b}}}{L{({1 + \frac{R_{b}R_{a}}{U_{b}U_{a}}})}})}}t}} \right)}},{t \geq 0}$ ${i_{a} = {\left( {1 + \frac{R_{b}u_{a}}{R_{a}u_{b}}} \right)\mspace{11mu} \left( {k_{b}\left( {1 - e^{{- \lambda}\; t}} \right)} \right)}},\; {t \geq 0}$

Initial and final conditions are:

${\begin{matrix} {{@t} = {{0\mspace{40mu} i_{b}} = 0}} & \; \\ {{@t} = {{0 + \frac{di}{dt}} = \frac{V_{S}}{L}}} & \; \\ {{@t} = {{\infty \mspace{31mu} i_{b}} = 0}} & \; \end{matrix}\therefore k_{b}} = {{\frac{V_{S}}{\left( {\frac{R_{a}}{u_{a}} + \frac{R_{b}}{u_{b}}} \right)}\mspace{14mu} {for}\mspace{14mu} R_{a}} = R_{b}}$ ${{and}\mspace{14mu} {finally}},{i_{a} = {\left( {1 + \frac{u_{a}}{u_{b}}} \right)\left( \frac{V_{S}}{\left( {\frac{R_{a}}{u_{a}} + \frac{R_{b}}{u_{b}}} \right)} \right)\mspace{11mu} \left( {1 - e^{{- \lambda}\; t}} \right)}}$

This result is similar to the result presented in the previous section yet with a slightly different form. Switch resistances are given by:

$R_{{SX}_{a}} = \frac{R_{a}}{u_{a}}$ $R_{{SX}_{b}} = \frac{R_{b}}{u_{b}}$

These resistances are time variant functions. They are also functions of T_(M).

It was shown in the previous section that unrestrained current in the switches produces an undesirable outphasing result if the goal is to mimic a classical outphasing transfer characteristic. The available reactive power in the first harmonic cycled through the inductor can be obtained from:

P _(X) _(L) =(0.707i _(a) ₁ )² x _(L)

x_(L) Δ Inductor Reactance

The first harmonic of the current was calculated previously from the Fourier analysis in the previous section and is given as:

$i_{a_{1}} = {A\left( {{a_{1}{\cos \left( \frac{2\; \pi \; t}{T_{C}} \right)}} + {b_{1}{\sin \left( \frac{2\; \pi \; t}{T_{C}} \right)}}} \right)}$

a₁, b₁ are the Fourier coefficients and A is an amplitude function associated with the gain of the differential equation, given above.

P_(X) _(L) can therefore be tailored to produce an appropriate transfer characteristic as a function of T_(M) by recognizing that i_(a), is a function of T_(M). Therefore, the following constraining equation is invoked:

${P_{X_{L}}\left( T_{M} \right)} = {{K\; {\cos^{2}\left( {\frac{2T_{M}}{T_{C}} \cdot \pi} \right)}} = {\left( {{.707} \cdot {i_{a_{1}}\left( T_{M} \right)}} \right)^{2}x_{L}}}$

This equation constrains the energy per unit time cycled through the inductor to vary as a function of T_(M) (outphasing delay time) according to the “ideal” classical outphasing transfer characteristic. The equation may be solved in terms of V_(S)(T_(M)), R_(SX) _(a) (T_(M)), R_(SX) _(b) (T_(M)), etc. That is, if the power source and/or the resistances are permitted a degree of freedom to vary as a function of T_(M), then the constraining equation possesses at least one solution.

These equations can be reduced considerably by rewriting and rearranging terms, using the local time span ramp approximation and associated Fourier analysis:

${A^{2}\left\{ {{R_{{SX}_{a}}\left( T_{M} \right)},{R_{{SX}_{b}}\left( T_{M} \right)}} \right\}} = \frac{K\; {\cos^{2}\left( {\frac{2T_{M}}{T_{C}} \cdot \pi} \right)}}{\left( {{{a_{1}\left( T_{M} \right)}\cos \frac{2\; \pi \; t}{T_{C}}} + {{b_{1}\left( T_{M} \right)}\sin \frac{2\; \pi \; t}{T_{C}}}} \right)^{2}}$

The left hand side is the portion which must be solved and is related to the Differential Equation Characteristic.

According to embodiments of the present invention, a numerical solution technique can be applied to obtain R_(SX) _(a) (T_(M)), R_(SX) _(b) (T_(M)). The denominator is a function of the energy of the ramped current pulses cycled in the inductor. a₁ and b₁ are the first harmonic terms of the Fourier component of these pulses. If the properties of the current pulse slope and amplitude can be controlled, then the available energy per unit time stored by the inductor can be tailored as well.

Examination of the differential equation for i_(a)(t) gives the following slope calculation:

${{{{{{{{{{{{\frac{d}{dt}i_{a}}}_{t = 0} = {{{E(t)} \cdot \frac{d}{dt}}\left( {1 - e^{{- \lambda}\; t}} \right)}}}_{t = 0} + {{\left( {1 - e^{{- \lambda}\; t}} \right) \cdot \frac{d}{dt}}{E(t)}}}}_{t = 0}\therefore{\frac{d}{dt}i_{a}}}}_{t = 0} = {\left( {1 + \frac{u_{a}R_{b}}{u_{b}R_{b}}} \right)\frac{V_{S}}{\left( {\frac{R_{a}}{u_{a}} + \frac{R_{b}}{u_{b}}} \right)}\lambda \; e^{{- \lambda}\; t}}}}_{t = 0} + {\frac{d}{dt}\left( {\left( {1 + \frac{u_{a}R_{b}}{u_{b}R_{b}}} \right)\left( \frac{V_{S}}{\frac{R_{a}}{u_{a}} + \frac{R_{b}}{u_{b}}} \right)\left( {1 - e^{{- \lambda}\; t}} \right)} \right)}}}_{t = 0}$ ${\frac{d}{dt}{i_{a}(t)}} = {{\frac{V_{S}}{L}@t} = 0}$

Hence, the slope is easy to calculate for the short duration of a single carrier cycle time. However, though the slope at t=0 is always constant, the slope at T_(C)/2<t<T_(C) can be modified by significant changes in the time constant τ_(eff)=R_(eff)/L, where R_(eff) is the effective resistance during the cycle. R_(eff) can change as a function of the desired modulation angle. Unfortunately, the calculation of the Fourier coefficients for the exponential pulse, though available in closed form, are very tedious expressions to write down and therefore will not be described. The coefficients a₁(T_(M)) and b₁(T_(M)) can be numerically evaluated.

FIGS. 50 and 51 illustrate how the pulse shape of the current changes as the switch resistance is permitted to vary significantly for increasing values of T_(M). In particular, FIG. 50 illustrates the case of fixed resistance in the switches of 0.1Ω. FIG. 51 provides a control which steps the resistance in 10Ω increments from 0.1Ω to 50.1Ω. As can be noted from FIGS. 50 and 51, energy cycling through the inductor decreases according to a true outphasing system when resistance is varied. The switch resistance or impedance increases with outphasing angle. Both duty cycle and current pulse amplitude vary. This blended control can be precisely tailored to obtain the ideal outphasing characteristic and in fact overcome any other non-linear or parasitic phenomena.

(c) Equations Related to Outphasing with a Load

The previous section dealt with available energy in an inductor cycled by outphased switches possessing internal real valued time varying impedances. This section examines a load circuit which is coupled by a capacitor. An example of such circuit is illustrated in FIG. 52.

The detailed transient solution is based on the solution of the following differential equation:

${{\frac{d^{2}}{{dt}^{2}}{il}} + \frac{\gamma_{2}}{\gamma_{1}} - {\frac{d}{dt}{il}} - {\frac{\gamma}{\gamma_{1}}{il}}} = {\frac{d}{dt}V_{S}}$ $\gamma_{0} = \left( {{\frac{R_{{SX}_{a}}}{R_{{SX}_{b}}}\left( {R_{{SX}_{b}} + R_{l}} \right)} + R_{l}} \right)$ $\gamma_{1} = {\frac{L}{R_{{SX}_{a}}}\gamma_{0}}$ $\gamma_{2} = {{\frac{L}{R_{{SX}_{a}}}\left( {\frac{1}{C}\left( {\frac{R_{{SX}_{a}}}{R_{{SX}_{b}}} + 1} \right)} \right)} - {\frac{R_{{SX}_{a}}}{R_{{SX}_{b}}}\left( {R_{{SX}_{b}} + R_{l}} \right)} + \gamma_{0}}$ $\gamma_{3} = {\frac{1}{C}\left( \frac{R_{{SX}_{a}}}{R_{{SX}_{b}}} \right)}$ $R_{{SX}_{a}} = \frac{R_{a}}{u_{a}}$ $R_{{SX}_{b}} = \frac{R_{b}}{u_{b}}$

u_(a) and u_(b) were defined previously as the cyclic unit step functions, which are also functions of the outphasing angle (as well as time).

Whenever R_(a) and R_(b) are constant, the differential equation can be solved in a piecewise manner for specific domains of time. The roots are calculated from:

$\beta_{1},{\beta_{2} = {\frac{- \gamma_{2}}{2\; \gamma_{1}} \pm \sqrt{{\frac{1}{4}\left( \frac{\gamma_{2}}{\gamma_{1}} \right)^{2}} - \frac{\gamma_{3}}{\gamma_{1}}}}}$

where R_(a), R_(b) are constant.

The current through the inductor can be written as:

$i_{a} = {{\frac{\gamma_{0}}{R_{{SX}_{a}}}{il}} + {\frac{d}{dt}\left( {\frac{1}{C}\left( {\frac{R_{{SX}_{a}}}{R_{{SX}_{b}}} + 1} \right)} \right)\; {il}}}$

The classical solution occurs whenever both switches are open, which yields:

$i_{l} = {{\frac{V_{S}}{L\left( {\alpha_{1} - \alpha_{2}} \right)}\left( {e^{{- \alpha_{2}}t}e^{{- \alpha_{1}}t}} \right)} + {{initial}\mspace{14mu} {conditions}}}$ $\alpha_{1},{\alpha_{2} = {\frac{- R_{l}}{2L} \pm \sqrt{{\left( \frac{R}{L} \right)^{2}\frac{1}{4}} - \frac{1}{LC}}}}$

Whenever the outphasing angles are small and both switches possess low values for R_(a), R_(b) then the previous analysis for cycling energy through the inductor applies for the times when switches are closed. When the switches open, the energy from the inductor is released and the load current becomes dominant as indicated by the classical solution.

The analysis is interesting when γ₀, γ₁, γ₂, γ are time variant. It is best to treat them as quasi-constant over a single charge and discharge cycle. This is reasonable if the values of R_(a) and R_(b) vary according to T_(M), the outphasing angle time delay. Fortunately, the envelope modulation rate is much less than the carrier rate for all practical scenarios of interest. With this, the first harmonics of the currents, i_(l) and i_(a) can be constrained such that;

$\frac{i_{l_{1}}^{2}\left( T_{M} \right)}{2R_{l}} = {A^{2}{\cos^{2}\left( {\varphi/2} \right)}}$

i_(l) ₁ ² Δ peak of the 1st harmonic (fundamental) flowing through R_(l)

This is the identical constraint placed on the inductor current given in the previous section. In that example, the load branch was non existent. The collapsing magnetic field of the inductor transfers energy in this example via the capacitor to R_(l). Furthermore, the RLC circuit sets up a resonator with Q. The Q of the circuit in the R-L circuit example is lowered to attain the transfer characteristic. As the resistances R_(a)(T_(M)), R_(b)(T_(M)) are increased to lower the Q on the charging cycle the Fourier coefficient a₁, b₁ are modified in i_(a) and i_(l) to lower the power transferred to the load.

FIG. 53 illustrates the currents through the inductor and the load in the example circuit of FIG. 52. FIG. 53 illustrates the cycles involved for storing energy in the inductor and the discharge into the load. Note that the current flowing in the load is filtered considerably by the tuned circuit.

When the switch source resistances vary, outphasing produces the desired attenuation effect for large angles of φ as illustrated by FIG. 54. FIG. 54 illustrates the effect of varying the switch source resistances on the currents through the inductor and the load in the example circuit of FIG. 52.

(i) Blended Outphasing Efficiency

This section provides a heuristic explanation for the efficiency performance of blended outphasing, as presented in sections 7 and 8.

Using blended outphasing, in every case where modified switching impedances are utilized, the minimum switch resistance is selected for the operating points requiring maximum efficiency. Furthermore, maximum efficiency is tailored according to the waveform envelope statistic. Typically, the probability of low power levels on the signal envelope is low. Therefore, the low efficiency circumstances for the circuit are minimal.

FIG. 56 illustrates a histogram associated with the WCDMA waveform. As shown, the greatest average probabilities are associated with slight back off from the peaks. At these high levels, the switch resistances are low and therefore possess relatively small T-R losses. The inductive tank has very high Q's for these low outphasing angles. Occasionally the envelopes make large excursions or even experience zero crossings. While efficiency drops as outphasing increases, these events occur infrequently.

Efficiency is calculated from the power in the 1^(st) harmonic dissipated by the load divided by the total battery power required to supply the circuit. The previous section developed the expressions for i_(l) and i₀ which permit the calculation. However, as pointed out in earlier analysis this calculation is not practical in closed form. Therefore, numerical analysis is typically preferred. Thus, efficiency is usually optimized numerically according to:

$\eta = \frac{\left( \frac{i_{l_{1}}^{2}}{2R_{l}} \right)}{\left( {{\overset{\sim}{i}}_{l_{a}} \cdot V_{S}} \right)}$

FIG. 57 illustrates the power output to the load as T_(M) varies in the example circuit of FIG. 52.

FIG. 58 illustrates the average DC current from battery in the example circuit of FIG. 52

11. SUMMARY

According to embodiments of the present invention, MISO amplifier/combiner circuits driven by VPA control algorithms outperform conventional outphasing amplifiers, including cascades of separate branch amplifiers and conventional power combiner technologies. Moreover, these circuits can be operated at enhanced efficiencies over the entire output power dynamic range by blending the control of the power source, source impedances, bias levels, outphasing, and branch amplitudes. These blending constituents are combined to provide an optimized transfer characteristic function, where optimization includes several aspects, including a well-behaved power transfer characteristic, overall efficiency on a per waveform basis, waveform specification performance such as enhanced EVM, ACPR, and unified architecture for all waveforms, i.e., universal waveform processor.

According to embodiments of the invention, VPA principles include the following.

-   -   a. Power is transferred from the power source to the load with         the greatest expediency. The number of levels of intermediate         processing between the power source and the load is minimized.     -   b. Consistent with a), power is not transferred through a power         combiner.     -   c. MISO inputs are controlled to direct the power flow from the         power source directly to the load, while modifying the spectral         content of the energy as it flows through the node, from the         power supply branch to the output or load branch.     -   d. Various controls are designed to optimize efficiency while         maintaining performance specifications of signals by operating         on the components of the complex envelope.     -   e. The physical structure of the MISO is significantly different         than traditional combiners in terms of impedance, frequency         response, and time delay. These properties permit the effective         application of the principles a) through d). Traditional         combiners cannot easily accommodate principles a), b) or c).

Simulation, mathematical analysis, and lab data agree and indicate that the MISO, when combined with the blended control algorithms, outperforms today's complex technologies and approaches efficiencies far greater than any available technology, while accommodating the ever increasing complex waveforms demanded by the cell phone market.

While various embodiments of the present invention have been described above, it should be understood that they have been presented by way of example only, and not limitation. It will be apparent to persons skilled in the relevant art that various changes in form and detail can be made therein without departing from the spirit and scope of the invention. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents. 

What is claimed is:
 1. A method for generating a radio frequency (RF) signal, the method comprising: generating, with a plurality of analog-to-digital converters, analog control signals; generating, with a modulator, a modulated control signal based on the analog control signals; generating, with an amplifier, a biased modulated control signal based on the modulated control signal and a first bias control signal; and generating the RF signal, with a multiple-input-single-output (MISO) device, based on the biased modulated control signal and a second bias control signal.
 2. The method of claim 1, further comprising: generating, with another modulator, another modulated control signal; and generating, with another amplifier, another biased modulated control signal based on the another modulated control signal and a third bias control signal.
 3. The method of claim 2, wherein the generating the RF signal comprises generating the RF signal based on the biased modulated control signal, the second bias control signal, and the another biased modulated control signal.
 4. The method of claim 1, wherein the generating the biased modulated control signal comprises controlling an input drive of the amplifier with the first bias control signal.
 5. The method of claim 1, wherein the generating the RF signal comprises controlling an input drive of the MISO device as a function of an envelope of the RF signal.
 6. The method of claim 5, wherein the input drive of the MISO device approaches zero when the envelope of the RF signal undergoes a change in phase.
 7. The method of claim 6, wherein the change in phase approaches 180 degrees.
 8. The method of claim 1, wherein the generating the RF signal comprises compensating for an asymmetry in at least one of an amplitude and a phase in the modulated control signal and the biased modulated control signal.
 9. The method of claim 1, wherein the generating the RF signal comprises concurrently adjusting the control signals, the first bias signal, and the second bias signal to change a phase in an envelope of the RF signal.
 10. The method of claim 1, wherein the RF signal conforms to the EDGE, GSM, CDMA2000, WCDMA, OFDM, or WLAN communication standard.
 11. An apparatus for generating a radio frequency (RF) signal, the apparatus comprising: a plurality of digital-to-analog converters (ADCs), wherein each of the ADCs is configured to generate an analog control signal; a modulator configured to generate a modulated control signal based on the analog control signals from the plurality of ADCs; an amplifier configured to generate a biased modulated control signal based on the modulated control signal and a first bias control signal and; a multiple-input-single-output (MISO) device configured to generate the RF signal based on the biased modulated control signal and a second bias control signal.
 12. The apparatus of claim 11, further comprising: another modulator configured to generate another modulated control signal; and another amplifier configured to generate another biased modulated control signal based on the another modulated control signal and a third bias control signal.
 13. The apparatus of claim 12, wherein the MISO device is configured to generate the RF signal based on the biased modulated control signal, the second bias control signal, and the another biased modulated control signal.
 14. The apparatus of claim 11, wherein the first bias control signal is configured to control an input drive of the amplifier.
 15. The apparatus of claim 11, wherein the second bias control signal is configured to control an input drive of the MISO device as a function of an envelope of the RF signal.
 16. The apparatus of claim 15, wherein the input drive of the MISO device approaches zero when the envelope of the RF signal undergoes a change in phase.
 17. The apparatus of claim 16, wherein the change in phase approaches 180 degrees.
 18. The apparatus of claim 11, wherein the first and second bias control signals are configured to compensate for an asymmetry in at least one of an amplitude and a phase in the modulated control signal and the biased modulated control signal.
 19. The apparatus of claim 11, wherein the control signals received by the ADCs, the first bias signal, and the second bias signal are adjusted concurrently to change a phase in an envelope of the RF signal.
 20. The apparatus of claim 11, wherein the RF signal conforms to the EDGE, GSM, CDMA2000, WCDMA, OFDM, or WLAN communication standard. 